View ISL6529_276279.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

TM
ISL6529
Data Sheet March 2002 FN9070
Dual Regulator-Synchronous-Rectified Buck PWM and Linear Power Controller
The ISL6529 provides the power control and protection for two output voltages in high-performance graphics cards and other embedded processor applications. The dual-output controller drives two N-Channel MOSFETs in a synchronous-rectified buck converter topology and one N-Channel MOSFET in a linear configuration. The ISL6529 provides both a regulated high current, low voltage supply and an independent, lower current supply integrated in an 14-lead SOIC package. The controller is ideal for graphic card applications where regulation of both the graphics processing unit (GPU) and memory supplies is required. The synchronous-rectified buck converter incorporates simple, single feedback loop, voltage-mode control with fast transient response. Both the switching regulator and linear regulator provide a maximum static regulation tolerance of 2% over line, load, and temperature ranges. Each output is user-adjustable by means of external resistors. An integrated soft-start feature brings both supplies into regulation in a controlled manner. Each output is monitored via the FB pins for undervoltage events. If either output drops below 51.5% of the nominal output level, both converters are shutdown.
Features
* Provides two regulated voltages - One synchronous-rectified buck PWM controller - One linear controller * Both controllers drive low cost N-Channel MOSFETs * 12V direct drive saves external components * Small converter size - 600kHz constant frequency operation - Small external component count * Excellent output voltage regulation - Both outputs: 2% over temperature * 5V down conversion * PWM and linear output voltage range: down to 0.8V * Simple single-loop voltage-mode PWM control design * Fast PWM converter transient response - High-bandwidth error amplifier - Full 0-100% duty ratio * Linear controller drives N-Channel MOSFET pass transistor * Fully-adjustable outputs * Undervoltage fault monitoring on both outputs
Ordering Information
PART NUMBER TEMP. RANGE ( oC) ISL6529CB ISL6529CB-T ISL6529EVAL1 0 to 70 PACKAGE 14 Ld SOIC PKG. NO. M14.15
Related Literature
* Technical Brief TB363 Guidelines for Handling and Processing Moisture Sensitive Surface Mount Devices (SMDs)
Pinout
ISL6529 (SOIC) TOP VIEW
LGATE 1 PGND 2 GND 3 5VCC 4 DRIVE2 5 FB2 6 NC 7 14 UGATE 13 12VCC 12 NC 11 NC 10 COMP 9 FB 8 NC
14Ld SOIC Tape and Reel Evaluation Board
Applications
* Graphics-GPU and memory supplies * ASIC power supplies * Embedded processor and I/O supplies * DSP supplies
NC = NO CONNECTION
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a trademark of Intersil Americas Inc. Copyright (c) Intersil Americas Inc. 2002. All Rights Reserved
Block Diagram
5VCC
VOLTAGE REFERENCE RESET (POR)
POWER-ON
0.80V
0.41V
1.28V
2
SHUTDOWN RESTART SOFTSTART AND FAULT LOGIC +5V INHIBIT SOFT-START +5VCC EA1 COMP1 OSCILLATOR PWM GATE LOGIC UV1 UV2 FB COMP
FB2
12VCC
12VCC
DRIVE2
UGATE
EA2
ISL6529
INHIBIT SOFT-START
LGATE
PGND
GND
ISL6529 Simplified Power System Diagram
+VIN +12V +5V
Q1 Q3 VOUT2 + LINEAR CONTROLLER PWM CONTROLLER +
VOUT1
Q2
ISL6529
Typical Application
+VIN (+5V or +3.3V) +5V CBP 5VCC C BP 12VCC +12V
CIN Q3 VOUT2 2.5V + FB2 DRIVE2
+
UGATE
Q1 PHASE
LOUT
VOUT1 1.5V +
COUT2
ISL6529
LGATE
Q2
COUT1
FB COMP
GND
PGND
3
ISL6529
Absolute Maximum Ratings
UGATE, LGATE, DRIVE2, . . . . . . . . . . . . . . . GND - 0.3V to 12VCC 5VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +7V 12VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +14V FB, FB2, COMP, . . . . . . . . . . . . . . . . . GND - 0.3V to 5VCC + 0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 4
Thermal Information
Thermal Resistance (Typical, Note 1) JA ( oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68 Maximum Junction Temperature (Plastic Package) . . . . . . . 150oC Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC (SOIC - Lead Tips Only)
Operating Conditions
Supply Voltage on 5VCC . . . . . . . . . . . . . . . . . . . . . . . . . +5V 10% Supply Voltage on 12VCC . . . . . . . . . . . . . . . . . . . . . . . +12V 10% Supply Voltage to drain of Upper MOSFETs . . . +3.3V to +5V 10% Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range. . . . . . . . . . . . . . . . . . . 0oC to 125oC
CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE: 1. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
PARAMETER VCC SUPPLY CURRENT Nominal Supply Current 12VCC Nominal Supply Current 5VCC POWER-ON RESET Rising 5VCC Threshold Falling 5VCCThreshold Rising 12VCC Threshold Falling 12VCCThreshold
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System Diagrams, and Typical Application Schematic SYMBOL TEST CONDITIONS MIN TYP MAX UNITS
ICC ICC
UGATE, LGATE and DRIVE2 Open UGATE, LGATE and DRIVE2 Open
-
2.7 3.5
3.0 4.5
mA mA
12VCC = 12V 12VCC = 12V 5VCC = 5V 5VCC = 5V
4.25 3.75 9.6 9.3
4.4 3.82 10.3 9.6
4.5 4.0 10.8 10.2
V V V V
OSCILLATOR AND SOFT-START Free Running Frequency Ramp Amplitude Soft-Start Interval REFERENCE VOLTAGE Reference Voltage System Accuracy PWM CONTROLLER ERROR AMPLIFIER DC Gain Gain-Bandwidth Product Slew Rate FB Input Current COMP High Output Voltage COMP Low Output Voltage COMP High Output, Source Current COMP Low Output, Sink Current Undervoltage Level (VFB/VREF) GBWP SR
FOSC DVOSC TSS
550 3.1
600 1.5 3.45
650 3.75
kHz V P-P ms
V REF
-2.0
0.800 -
+2.0
V %
RL = 10k, CL = 10pf RL = 10k, CL = 10pf RL = 10k, CL = 10pf VFB = 0.8V
3.0 2.5 3.5 -
80 15 6 20 4.5 0.5 3.3 5.0 51.5
150 1.0 -
dB MHz V/s nA V V A mA %
II
VOUT High VOUT Low IOUT High IOUT Low VUV
4
ISL6529
Electrical Specifications
PARAMETER PWM CONTROLLER GATE DRIVERS UGATE and LGATE Maximum Voltage UGATE and LGATE Minimum Voltage UGATE and LGATE Source Current UGATE and LGATE Sink Current UGATE and LGATE OUTPUT IMPEDANCE LINEAR REGULATOR (DRIVE2) DC Gain Gain-Bandwidth Product Slew Rate FB2 Input Current Drive2 High Output Voltage Drive2 Low Output Voltage Drive2 High Output Source Current Drive2 Low Output Sink Current Over-Voltage Level (VFB2/VREF) Under-Voltage Level (VFB2/VREF) REGULATOR ISOLATION Change in Linear Regulator Output Voltage2 Change in PWM Regulator Output Voltage2 2. Measured in the evaluation board. Vout Vout Linear Output = 2.5V, 6A Load Change on PWM PWM Output = 1.5V, 1A Load Change on Linear <0.5 <0.5 % % GBWP SR
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System Diagrams, and Typical Application Schematic (Continued) SYMBOL TEST CONDITIONS MIN TYP MAX UNITS
VHGATE VLGATE IGATE IGATE RDS(on)
12VCC = 12V 12VCC = 12V 12VCC = 12V 12VCC = 12V 12VCC = 12V
11 -
12 0 -1 1 3.1
0.5 4.3
V V A A
RL = 10k, CL = 10pf RL = 10k, CL = 10pf RL = 10k, CL = 10pf VFB2 = 0.8V
9.5 -0.7 0.85
80 15 6 20 10.3 0.1 -1.4 1.2 160 51.5
150
dB MHz V/s nA V
II
VOUT High VOUT Low IOUT High IOUT Low VOV VUV Percent of Nominal Percent of Nominal
1.0 -
V A mA % %
-
Functional Pin Descriptions
LGATE 1 PGND 2 GND 3 5VCC 4 DRIVE2 5 FB2 6 NC 7 14 UGATE 13 12VCC 12 NC 11 NC 10 COMP 9 FB 8 NC NC = NO CONNECTION
5VCC (Pin 4)
Provide a well decoupled 5V bias supply for the IC to this pin. The voltage at this pin is monitored for Power-On Reset (POR) purposes.
DRIVE2(Pin 5)
Connect this pin to the gate terminal of an external N-Channel MOSFET transistor. This pin provides the gate voltage for the linear regulator pass transistor. It also provides a means of compensating the error amplifier for applications where the user needs to optimumize the regulator transient response.
LGATE (Pin 1)
Lower gate drive output. Connect to gate of the low-side MOSFET.
FB2 (Pin 6)
Connect the output of the linear regulator to this pin through a properly sized resistor divider. The voltage at this pin is regulated to 0.8V. This pin is also monitored for undervoltage events. Pulling and holding FB2 above 1.28V shuts down both regulators. Releasing FB2 initiates soft-start on both regulators.
PGND (Pin 2)
This pin is the power ground return for the lower gate driver.
GND (Pin 3)
Signal ground for the IC. All voltage levels are measured with respect to this pin. Place via close to pin to minimize impedance path to ground plane.
NC (Pins 7, 8, 11, and 12)
No internal connection.
5
ISL6529
FB (Pin 9) and COMP (Pin 10)
FB and COMP are the available external pins of the error amplifier. The FB pin is the inverting input of the error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-mode control feedback loop of the standard synchronous-rectified buck converter. voltage. As the error amplifier voltage increases, the pulsewidth on the UGATE pin increases to reach its steady-state duty cycle at time t2. The error amplifier reference of the linear controller also rises relative to the soft-start reference. The resulting soft ramp on DRIVE2 brings VOUT2 within regulation limits by time t2.
12VCC(Pin 13)
Provides bias voltage for the gate drivers.The voltage at this pin is monitored for Power-On Reset (POR) purposes.
+5V (VCC)
UGATE(Pin 14)
Connect UGATE to the upper MOSFET gate. This pin provides the gate drive for the MOSFET.
+3.3V (UPPER FET DRAIN)
Description
Operation Overview
The ISL6529 monitors and precisely controls two output voltage levels. Refer to the Block Diagram, Simplified Power System Diagram, and Typical Application Schematic on pp. 2-3. The controller is intended for use in graphics cards or embedded processor applications with 5V and 12V bias input available. The IC integrates both a synchronous-rectified buck PWM controller and a linear controller. The PWM controller is designed to regulate the high current GPU voltage (V OUT1). The PWM controller regulates the output voltage to a level programmed by a resistor divider. The linear controller is designed to regulate the lower current local memory voltage (VOUT2 ) through an external N-Channel MOS pass transistor.
0V (1V/DIV) VOUT2 (2.5V)
VOUT1 (1.5V)
0V
(0.5V/DIV) t0 t1 t2 TIME
Initialization
The ISL6529 automatically initializes upon application of input power. Special sequencing of the input supplies is not necessary. The POR function continually monitors the input bias supply voltage at the 5VCC and 12VCC pins. The POR function initiates soft-start operation after these supply voltages exceed their POR threshold voltages.
FIGURE 1. SOFT-START INTERVAL
Undervoltage Protection
The FB and FB2 pins are monitored during converter operation by two separate undervoltage (UV) comparators. If the FB voltage drops below 51.5% of the reference voltage (0.41V), a fault signal is generated. The internal fault logic shuts down both regulators simultaneously when the fault signal triggers a restart. Figure 2 (next page) illustrates the protection feature responding to an UV event on VOUT1. At time t0, VOUT1 has dropped below 51.5% of the nominal output voltage. Both outputs are quickly shut down and the internal soft-start function begins producing soft-start ramps. The delay interval, t0 to t3, seen by the output is equivalent to three soft-start cycles. After a short delay interval of 10.5ms, the fourth internal soft-start cycle initiates a normal soft-start ramp of the output, at time t3. Both outputs are brought back into regulation by time t4, as long as the UV event has cleared. Had the cause of the UV still been present after the delay interval, the UV protection circuitry becomes active approximately 875ms into the soft-start interval. A fault
Soft-Start
The POR function initiates the digital soft-start sequence. Both the linear regulator error amplifier and PWM error amplifier reference inputs are forced to track a voltage level proportional to the soft-start voltage. As the soft-start voltage slews up, the PWM comparator regulates the output relative to the tracked soft-start voltage, slowly charging the output capacitor(s). Simultaneously, the linear output follows the smooth ramp of the soft-start function into normal regulation. Figure 1 shows the soft-start sequence for a typical application. At t0, the 5VCC and 12VCC bias voltages start to ramp followed by the 3.3V input supply. Once the voltage on 5VCC and 12VCC cross the POR thresholds at time t1, both outputs begin their soft-start sequence. The triangle waveform from the PWM oscillator is compared to the rising error amplifier output
6
ISL6529
signal could then be generated and the outputs once again shutdown. The resulting hiccup mode style of protection would continue to repeat indefinitely. However, since the value of R1 affects the values of the rest of the compensation components, it is advisable to keep its value less than 5k. Depending on the value chosen for R1, R4 can be calculated based on the following equation:
R1 x 0.8V R4 = ------------------------------------VOUT1 - 0.8 V
VOUT2 (2.5V)
(EQ. 1)
If the output voltage desired is 0.8V, simply route VOUT1 back to the FB pin through R1, but do not populate R4.
VOUT1 (1.5V)
DELAY INTERVAL 0V (0.5V/DIV) VOUT2 (2.5V)
The linear regulator output voltage is also set by means of an external resistor divider as shown in Figure 4. The two resistors used to set the output voltage should not exceed a parallel equivalent value, referred to as RFB, of 5k. This restriction is due to the manner of implementation of the softstart function. The following relationship must be met:
R5 x R6 R FB = --------------------- < 5k R5 + R6
(EQ. 2)
INTERNAL SOFT-START FUNCTION +3.3VIN DELAY INTERVAL VOUT2 + COUT2 R5 R6 Q3 R12 DRIVE2 C4 FB2
0V t0 t1 t2 TIME t3 t4
ISL6529
FIGURE 2. UNDERVOLTAGE PROTECTION RESPONSE
Output Voltage Selection
The output voltage of the PWM converter can be programmed to any level between VIN (i.e. +3.3V) and the internal reference, 0.8V. An external resistor divider is used to scale the output voltage relative to the reference voltage and feed it back to the inverting input of the error amplifier, see Figure 3.
+12V +5V
FIGURE 4. OUTPUT VOLTAGE SELECTION OF THE LINEAR
To ensure the parallel combination of the feedback resistors meets this criteria, choose a target value for R FB of less than 5k and then apply the following equations:
V OUT 2 R5 = ------------------ x R FB VREF R5 x V REF R6 = ---------------------------------------V OU T2 - V REF
12VCC 5VCC +3.3V
VOUT1
+
LOUT
Q1
UGATE
ISL6529
Q2 LGATE
COUT1
where VOUT2 is the desired linear regulator output voltage and VREF is the internal reference voltage, 0.8V. For an output voltage of 0.8V, simply populate R5 with a value less than 5kW and do not populate R6.
FB R3 R1 C3 R2 R4 C1 C2 COMP
Converter Shutdown
Pulling and holding the FB2 pin above a typical threshold of 1.28V will shutdown both regulators. Upon release of the FB2 pin, the regulators enter into a soft-start cycle which brings both outputs back into regulation.
FIGURE 3. OUTPUT VOLTAGE SELECTION OF THE PWM
7

R5 V OU T2 = 0.8 x 1 + ------R6

(EQ. 3)
(EQ. 4)
ISL6529 PWM Controller Feedback Compensation
A simplified representation of the voltage-mode control loop used for output regulation by the converter is shown in Figure 5. The output voltage, V OUT, is fed back to the negative input of the error amplifier which is regulated to the reference voltage level, VREF. The error amplifier output, VE/A, is compared with the triangle wave produced by the oscillator, VOSC, to provide a pulse-width modulated (PWM) signal from the PWM comparator. This signal is then used to switch the MOSFET and produce a PWM waveform with an amplitude of VIN at the PHASE node. The square-wave PHASE voltage is then smoothed by the output filter, LOUT and COUT, to produce a DC voltage level. The modulator transfer function is defined as VOUT/VE/A . The internal PWM comparator and driver circuits equate to a DC gain block dominated by the supply voltage, VIN , divided by the peak-to-peak magnitude of the triangle wave, VOSC. The output filter components, LOUT and COUT, shape the overall modulator small-signal transfer function by contributing a double pole break frequency at FLC and a zero at FESR .
VIN OSC PWM COMP DRIVER LOUT PHASE CO +
Modulator Break Frequency Equations
1 FLC = --------------------------------------2 x L O x C O 1 F ESR = ---------------------------------------2 x ESR x C O
(EQ. 5) (EQ. 6)
The compensation network consists of the error amplifier and the impedance networks ZIN and ZFB . They provide the link between the modulator transfer function and a controllable closed loop transfer function of VOUT/VREF. The goal of component selection for the compensation network is to provide a loop gain with high 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees.
Compensation Break Frequency Equations
Poles:
1 FP1 = -----------------------------------------------------C1 x C2 2 x R 2 x --------------------C1 + C2 1 FP2 = ----------------------------------2 x R 3 x C3
(EQ. 7)
Zeros:
1 F Z1 = ----------------------------------2 x R 2 x C1
VOUT
VOSC
-
+
1 F Z2 = -----------------------------------------------------2 x ( R1 + R3 ) x C3
ZFB VE/A ZIN VREF
ESR (PARASITIC)
Follow this procedure for selecting compensation components by locating the poles and zeros of the compensation network: 1. Set the loop gain (R2/R1) to provide a converter bandwidth of one quarter of the switching frequency. 2. Place the first compensation zero, FZ1, below the output filter double pole (~75% FLC). 3. Position the second compensation zero, FZ2, at the output filter double pole, FLC. 4. Locate the first compensation pole, FP1, at the output filter ESR zero, FESR. 5. Position the second compensation pole at half the converter switching frequency, FSW. 6. Check gain against error amplifier's open-loop gain. 7. Estimate phase margin; repeat if necessary.
+ ERROR AMP
DETAILED COMPENSATION COMPONENTS C2 C1 R2 ZFB ZIN C3 R1 R3 VOUT
COMP
+
FB
ISL6529
0.8V
FIGURE 5. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN
8

(EQ. 8)
(EQ. 9) (EQ. 10)
ISL6529
FZ1 100 80 60 GAIN (dB) 40 20 0 -20 -40 -60 FZ2 FP1 FP2 OPEN LOOP ERROR AMP GAIN

Application Guidelines
Layout Considerations
Layout is very important in high frequency switching converter design. With power devices switching efficiently at 600kHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device overvoltage stress. Careful component layout and printed circuit board design minimizes the voltage spikes in the converters. As an example, consider the turn-off transition of the PWM MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the lower MOSFET and parasitic diode. Any parasitic inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimizes the magnitude of voltage spikes. There are two sets of critical components in a DC-DC converter using the ISL6529. The switching components are the most critical because they switch large amounts of energy, and therefore tend to generate large amounts of noise. Next are the small signal components which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 7 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections through vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminal to the output inductor short. The power plane should support the input and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase node. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the UGATE pin to the MOSFET gate should be kept short and wide enough to easily handle the 1A of drive current.
V IN 20 log ----------------V OSC

COMPENSATION GAIN
MODULATOR GAIN 10 100
FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Figure 6 shows an asymptotic plot of the DC-DC converter's gain vs. frequency. The actual modulator gain has a high gain peak dependent on the quality factor (Q) of the output filter, which is not shown in Figure 6. Using the above procedure should yield a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin.
Linear Regulator Feedback Compensation
The regulator may be compensated with a series 6.8k resistor and a 470pF capacitor connected between FB2 and DRIVE2. This will provide compensation for all loads and ranges of output capacitor values and a range of capacitor ESR values from aluminum electrolytic to low-ESR organic polymer capacitors. This will not insure optimum load transient response since the regulator system, like an internally compensated operational amplifier is overcompensated. To optimize transient response, when required, the regulator should be in the actual application circuit with the desired output capacitors and associated PC board parasitics and load. The value of C4 would be reduced and the series resistor, R12 adjusted for optimum rise and fall time, with a minimum of overshoot.

R2 20 log -------R1

FLC 1K
FESR 10K 100K 1M
LOOP GAIN
10M
FREQUENCY (Hz)
9
ISL6529
The switching components should be placed close to the ISL6529 first. Minimize the length of the connections between the input capacitors, C IN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper MOSFET and lower diode and the load.
+3.3 VIN +5 VCC 5VCC GND C BP CIN
Component Selection Guidelines
Output Capacitor Selection
Output capacitors are required to filter the output and supply the load transient current. The filtering requirements are a function of switching frequency and output current ripple. The load transient requirements are a function of the transient load current slew rate (di/dt) and magnitude. These requirements are generally met with a mix of capacitors and careful layout.
PWM Regulator Output Capacitors
Modern digital ICs can produce high transient load slew rates. High frequency capacitors initially supply the transient current and slow the load rate-of-change seen by the bulk capacitors. The bulk filter capacitor selection is generally determined by the effective series resistance (ESR) and voltage rating requirements rather than actual capacitance requirements.
VOUT1 LOAD
+12 VCC 12VCC PGND UGATE CBP Q1 PHASE Q2 C2 R2 FB R4 LOUT
ISL6529
LGATE COMP
COUT1
C1 R1 C3 R3 +3.3 VIN Q3 R5 VOUT2 LOAD
High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Specialized low-ESR capacitors intended for switchingregulator applications are recommended for the bulk capacitors. The bulk capacitor's ESR determines the output ripple voltage and the initial voltage drop following a high slew-rate transient edge. Aluminum electrolytic, tantalum, and special polymer capacitor ESR values are related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor's impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor.
DRIVE2 FB2 R6 KEY
C OUT2
ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE
FIGURE 7. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
The critical small signal components include any bypass capacitors, feedback components, and compensation components. Position the bypass capacitors, C BP, close to the VCC pin with a via directly to the ground plane. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors for both regulators should also be located as close as possible to the relevant FB pin with vias tied straight to the ground plane as required.
PWM Output Inductor Selection
The PWM converter requires an output inductor. The output inductor is selected to meet the output voltage ripple requirements and sets the converter response time to a load transient. The inductor value determines the converter's ripple current and the ripple voltage is also a function of the ripple current. The ripple voltage and current are approximated by the following equations:
V IN - V OUT V OU T I = ------------------------------- x --------------V IN FS x L V OU T = I x ESR
(EQ. 11)
(EQ. 12)
10
ISL6529
Increasing the value of inductance reduces the output ripple current and voltage ripple. However, increasing the inductance value will slow the converter response time to a load transient. One of the parameters limiting the converter's response to a load transient is the time required to slew the inductor current. Given a sufficiently fast control loop design, the ISL6529 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time interval required to slew the inductor current from an initial current value to the final current level. During this interval the difference between the inductor current and the load current must be supplied by the output capacitor(s). Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load:
L O x ITRAN tRI SE = ------------------------------V IN - V OUT L O x ITRAN t FALL = -----------------------------VOUT
For a through-hole design, several aluminum electrolytic capacitors may be needed. For surface mount designs, tantalum or special polymer capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. TRANSISTOR SELECTION/CONSIDERATIONS The ISL6529 requires three external transistors. One N-Channel MOSFET is used as the upper switch in a standard buck topology PWM converter. Another MOSFET is used as the lower synchronous switch. The linear controller drives the gate of an N-Channel MOS transistor used as the series pass element. The MOSFET transistors should be selected based upon rDS(ON) , gate supply requirements, and thermal management considerations.
Upper MOSFET SWITCH Selection
In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses account for a large portion of the power dissipation of the upper MOSFET. Switching losses also contribute to the overall MOSFET power loss.
2 PConductionUpper I o x r DS ( on ) x D
(EQ. 13)
(EQ. 14)
(EQ. 15) (EQ. 16)
where ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +3.3V input source, the worst case response time can be either at the application or removal of load and dependent upon the output voltage setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time.
1 PSwitching -- I o x VIN x tSW x F SW 2
where Io is the maximum load current, D is the duty cycle of the converter (defined as V O/VIN ), tSW is the switching interval, and FSW is the PWM switching frequency. The lower MOSFET has only conduction loses since it switches with zero voltage across the device. Conduction loss is:
2 PConductionLow er I o x r DS ( on ) x ( 1 - D ) (EQ. 17)
Input Capacitor Selection
The important parameters for the bulk input capacitors are the voltage rating and the RMS current rating. For reliable operation, select bulk input capacitors with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 of the summation of the DC load current. Use a mix of input bypass capacitors to control the voltage overshoot across the switching MOSFETs. Use ceramic capacitance for the high frequency decoupling and bulk capacitors to supply the RMS current. Small ceramic capacitors can be placed very close to the upper MOSFET to suppress the voltage induced in the parasitic circuit impedances. Connect them directly to ground with a via placed very close to the ceramic capacitor footprint. 11
These equations assume linear voltage-current transitions and are approximations. The gate-charge losses are dissipated by the ISL6529 and do not heat the MOSFET. However, large gate-charge increases the switching interval, tSW, which increases the upper MOSFET switching losses. Ensure that the MOSFET is within its maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature, air flow, and load current requirements. The gate drive to the switching transistors ranges from slightly below 12V to ground. Because of the large voltage swing, logic-level transistors are not necessary in this application.
ISL6529
However, if logic-level transistors or transistors with low VGS(on) are used, close attention to layout guidelines should be exercised, as the low gate threshold could lead to some shoot-through despite counteracting circuitry present aboard the ISL6529. The power dissipated in the linear regulator is:
PLI NEAR IO x ( V I N - V OU T )
(EQ. 18)
N-Channel MOSFET Transistor Selection
The main criteria for selection of the linear regulator pass transistor is package selection for efficient removal of heat. Select a package and heatsink that maintains the junction temperature below the rating with a maximum expected ambient temperature.
where IO is the maximum output current and VOUT is the nominal output voltage of the linear regulator.
Linear Regulator Output Capacitors
The output capacitors for the linear regulator provide dynamic load current. Output capacitors should be selected for transient load regulation.
ISL6529 Converter Application Circuit
L2 +3.3V C7 1000F 1H C6 470F C8 4.7F
+5V
+12V
C5 1F
C15 1F VOUT1 (6A) 1.5V
5VCC Q3 VOUT2 (1A) 2.5V C14 1F C4 470pF R12 6.8k DRIVE2
12VCC
UGATE LGATE COMP
Q1 Q2
L1 4.7H C9 470F C10 1F
ISL6529
R5 4.64k C12 1500F
FB2 GND
R6 2.15k
FB PGND
C1 47nF R2 10.7k
C2 33pF R1 1k
Q1, Q2 Q3 L1 L2
IRF7313 MTD3055V 919AS-4R7M 919AS-1R0N
IR Fairchild TOKO TOKO
R4 1.15k
R3 10
C3 47nF
FIGURE 8. POWER SUPPLY APPLICATION CIRCUIT FOR A GRAPHICS CONTROLLER
12
ISL6529 Small Outline Plastic Packages (SOIC)
N INDEX AREA H E -B1 2 3 SEATING PLANE -AD -CA h x 45o 0.25(0.010) M BM
M14.15 (JEDEC MS-012-AB ISSUE C)
14 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE INCHES SYMBOL A
L
MILLIMETERS MIN 1.35 0.10 0.33 0.19 8.55 3.80 5.80 0.25 0.40 14 0o MAX 1.75 0.25 0.51 0.25 8.75 4.00 6.20 0.50 1.27 8o NOTES 9 3 4 5 6 7 Rev. 0 12/93
MIN 0.0532 0.0040 0.013 0.0075 0.3367 0.1497 0.2284 0.0099 0.016 14 0o
MAX 0.0688 0.0098 0.020 0.0098 0.3444 0.1574 0.2440 0.0196 0.050 8o
A1 B C D E e
C
e
B 0.25(0.010) M C AM BS

A1 0.10(0.004)
0.050 BSC
1.27 BSC
H h L N
NOTES: 1. Symbols are defined in the "MO Series Symbol List" in Section 2.2 of Publication Number 95. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension "D" does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension "E" does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. "L" is the length of terminal for soldering to a substrate. 7. "N" is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width "B", as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation's quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
Sales Office Headquarters
NORTH AMERICA Intersil Corporation 7585 Irvine Center Drive Suite 100 Irvine, CA 92618 TEL: (949) 341-7000 FAX: (949) 341-7123 Intersil Corporation 2401 Palm Bay Rd. Palm Bay, FL 32905 TEL: (321) 724-7000 FAX: (321) 724-7946 EUROPE Intersil Europe Sarl Ave. William Graisse, 3 1006 Lausanne Switzerland TEL: +41 21 6140560 FAX: +41 21 6140579 ASIA Intersil Corporation Unit 1804 18/F Guangdong Water Building 83 Austin Road TST, Kowloon Hong Kong TEL: +852 2723 6339 FAX: +852 2730 1433
13

▲Up To Search▲

Price & Availability of ISL6529

	To Download ISL6529 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .